1. Field of the Invention
The present invention relates to a wireless communication system that implements wireless communication by adaptively controlling radio beams, and a method of controlling the same.
2. Description of Related Art
Firstly, documents will be referred to in this specification are listed below.    Document 1: International Patent Publication WO/2008/090836    Document 2: Japanese Unexamined Patent Application Publication No. 2006-245986.    Document 3: Japanese Unexamined Patent Application Publication No. 2000-165959.    Document 4: United States Patent Publication No. 2007/0205943.    Document 5: K. Maruhashi et al., “60-GHz-band LTCC Module Technology for Wireless Gigabit Transceiver Applications”, IEEE International Workshop on Radio-Frequency Integration Technology, Digest, pp. 131-134, December 2005.    Document 6: K. Ohata et al., “1.25 Gbps wireless Gigabit Ethernet Link at 60-GHz-Band”, IEEE MTT-S International Microwave Symposium, Digest, pp. 373-376, June 2003.    Document 7: J. F. Buckwalter et al., “An Injected Subharmonic Coupled-Oscillator Scheme for a 60-GHz Phased-Array Transmitter”, IEEE Transactions on Microwave Theory and Techniques, Vol. 12, pp. 4271-4280, December 2006.    Document 8: S. Alausi et al., “A 60 GHz Phased Array in CMOS”, IEEE 2006 Custom Integrated Circuits Conference, Digest, pp. 393-396, San Jose, September 2006.    Document 9: J. Capon, “High-Resolution Frequency-Wavenumber Spectrum Analysis”, Proceedings of the IEEE, Vol. 57, No. 8, pp. 1408-1418, August 1969.    Document 10: K. Kumaresan et al., “Estimating the angles of arrival of multiple plane waves”, IEEE Transactions on Aerosp. Electron. Syst., Vol. AES-19, pp. 134-139, January 1983.    Document 11: P. Stoica et al., “MUSIC, Maximum Likelihood, and Cramer-Rao Bound”, IEEE Transactions on Acoustics, Speech, and Signal Processing, Vol. 37, No. 5, pp. 720-741, May 1989.    Document 12: R. Roy et al., “ESPRIT-Estimation of Signal Parameters Via Rotational Invariance Techniques”, IEEE Transactions on Acoustics, Speech, and Signal Processing, Vol. 37, No. 7, pp. 984-995, July 1989.    Document 13: I. Lakkis et al., “IEEE P802.15 Working Group for Wireless Personal Area Networks (WPANS): TG3c Call for Proposals”, 15-08-0355-00-003c, May 2008.    Document 14: K. Sato et al., “Channel model for millimeter-wave WPAN”, The 18th Annual IEEE International Symposium on Personal, Indoor and Mobile Radio communications (PIMRC '07), 2007.    Document 15: “Propagation data and prediction methods for the planning of indoor radio communication systems and radio local area networks in the frequency range 900 MHz to 100 GHz,” ITU-R, P. 1238-3, April, 2003
In recent years, use of wireless communication devices using wideband millimeter waves (30 GHz to 300 GHz) has become increasingly widespread. The millimeter-wave radio technology has been expected to become applicable, especially, to high-rate radio data communication in the order of gigabit, such as radio transmission of high-resolution images (for example, see Documents 5, 6 and 7).
Millimeter waves, which have high frequencies, have a high rectilinear propagation property, and therefore it poses a problem in a case where radio transmission is to be implemented indoors. In addition to the high rectilinear propagation property, the millimeter waves are also significantly attenuated by a human body or a similar object. Therefore, if a person stands between the transmitter and the receiver in a room or the like, it cannot get an unobstructed view, thus making the transmission very difficult (the shadowing problem). Since this problem results from the fact that the higher rectilinear propagation property of radio waves has become higher with increase in frequency and therefore the propagation environment has been changed, it is not limited to the millimeter waveband (30 GHz and above). Although it is not easy to clearly specify the transition frequency, it has been believed to be around 10 GHz. Meanwhile, according to recommendations of the International Telecommunications Union (Document 15), a power loss coefficient, which indicates the attenuation amount of a radio wave with respect to the propagation distance, is 22 for 60 GHz in an office, while it is 28 to 32 for 0.9 to 5.2 GHz, in offices. Considering that it is 20 in the case of free-space loss, the effects of scattering and diffraction and the like are considered to be small for higher frequencies in the order of 60 GHz.
To solve the problem described above, Document 2, for example, describes a system in which a plurality of transmission paths are provided by installing a plurality of receiving units in the receiving device, so that when one of the transmission paths between the transmitting device and the receiving units is shielded, the transmission is carried out with another transmission path. Furthermore, as another method for solving the problem, Document 3 describes a contrivance to secure a plurality of transmission paths by installing reflectors on a ceiling and walls.
In the method described in Document 2, it is very difficult to continue the communication when shielding occurs in the vicinity of the transmitting device or when all of the installed receiving units are shielded. Meanwhile, the method described in Document 3 requires users to consider particular conditions such as a condition that reflectors need to be installed with taking the positions of the transmitter and the receiver into account.
However, recent studies on propagation properties of millimeter waves have found out that reflected waves could be utilized without intentionally installing reflectors. FIG. 16 shows a configuration of a system using a wide-angle antenna. FIG. 17 shows an example of a delay profile of a system using a wide-angle antenna like the one shown in FIG. 16 when the system is used indoors. In the system using a wide-angle antenna like the one shown in FIG. 16, the received power is maximized when the main wave, which is arrives faster than any other waves, arrives at the receiver as shown in FIG. 17. After that, although delayed waves, such as second and third waves, also arrive, the received power of these waves is smaller than that of the main wave. These second and third waves are waves reflected from the ceiling and the walls. This situation is remarkably different from the propagation environment of radio waves having a lower rectilinear propagation property, such as a 2.4 GHz band used in wireless LANs (Local Area Networks). In the 2.4 GHz band, it is very difficult to clearly separate waves in their incoming directions (DoAs: Directions of Arrival) because of the effects of diffraction and multiple reflections. On the other hand, in the millimeter waves having a high rectilinear propagation property, although radio waves are relatively clearly distinguished in their incoming directions, the number of delayed waves is limited and their received-signal levels are small.
Therefore, when the direct wave (main wave) is shielded, it is necessary to ensure a sufficient received-signal level at the receiver by pointing a narrow beam having a high directive gain to the incoming direction of a reflected wave as shown in FIG. 15 in order to continue the transmission by using the reflected wave. However, in order to eliminate the necessity for users to consider the particular conditions such as a condition for the relative positions of the transmitter and receiver, a beam forming technique capable of dynamically controlling the direction of a narrow beam is indispensable.
In the beam forming, it is necessary to construct an antenna array. For millimeter waves having a short wavelength (e.g., 5 mm in the case of a frequency of 60 GHz), the antenna array can be implemented in a small area. Phase shifter arrays and oscillator arrays for use in such antenna arrays for millimeter waves have been developed (for example, see Documents 7 and 8).
Furthermore, a DoA (Direction of Arrival) estimation technique has been known as a technique for a different purpose from the beam forming using an antenna array. The DoA estimation technique is a technique for use in, for example, radars, sonar, and propagation environment measurements, and used for estimating the incoming directions and the power of radio waves to be received at antenna arrays with high accuracy. Various methods including a beam former method and Capon method (Document 9), a linear estimation method, a minimum norm method (Document 10), MUSIC (Multiple Signal Classification) (Document 11), and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques) (Document 12) have been known as an algorithm used for such techniques.
When this DoA estimation technique is used in propagation environment measurement with an installed radio wave source, an omni-antenna (nondirectional antenna) is often used as the radio wave source. Document 14 shows an example of such a technique.
The present inventors have found that, when the direct wave is shielded and the radio transmission is to be continued by using a reflected wave in indoor millimeter-wave systems, a following problem arises.
When the wave (direct wave, reflected wave) that is actually used is switched, it is desirable to minimize the time during which the transmission is disconnected (hereinafter called “transmission-disconnected time”). Such minimization of the transmission-disconnected time becomes especially an important requirement, for example, in the transmission of non-compressed images for which real-time capability is required. Meanwhile, when a reflected wave is used, it is necessary to increase the directive gain of the antenna, and thereby to increase the reception strength, by narrowing the antenna beam width.
However, the directions (steps) in which the searches need to be carried out increases as the beam width becomes narrower. Therefore, the time necessary to find and establish the beam direction with which the incoming wave is effectively received becomes longer, and therefore the transmission-disconnected time also becomes longer. Accordingly, it has been desired to develop a beam direction setting method that can shorten the transmission-disconnected time even in such situations. It should be noted that even the use of an apparatus capable of temporally storing data is impractical because a huge buffer memory is required when the transmission-disconnected time becomes longer.
FIG. 4 shows a configuration example of a transmitting/receiving device used in the beam forming. Note that circuits unnecessary for the explanation of the operation are omitted in the figure. It has M transmitting antennas and N receiving antennas. A transmitter 401 includes a transmitter circuit 403 to which external data is input. The output of the transmitter circuit 403 is branched into M signals, and they are input to the respective AWV (array weight vector) control circuits 404-1-404-M. Each signal is changed either in its amplitude or in its phase, or both in its amplitude and in its phase in the corresponding AWV control circuit, and finally output through a transmitting antenna array composed of respective antenna elements 405-1-405-M. Each of the AWV control circuits 404-1-404-M can be implemented by, for example, the series connection of an analog phase shifter and a variable-gain amplifier. In such a configuration, both the amplitude and phase of a signal may be controlled in a continuous manner. Furthermore, when the AWV control circuits 404-1-404-M are implemented by digital phase shifters, only the phases of signals are controlled in a discrete manner. AWVs that are controlled by the AWV control circuits 404-1-404-M are, in general, expressed as the following expression (1):{right arrow over (W)}≡[w1,w2, . . . ,wM]T  (1)where w1, w2, . . . wM are complex numbers and the superscript T indicates transposition. Furthermore, when only the phases are controlled, the expression (1) can be expressed as the following expression (2):{right arrow over (W)}≡[ejθ1,ejθ2, . . . ,ejθM]T  (2)where θ1, θ2, . . . , θM are phase control amounts.
Furthermore, a process/arithmetic circuit 406 provides instructions on the AWV setting of the AWV control circuits 404-1-404-M through a control circuit 407. With the change in both of the amplitude and phase or either one of them that is made to each signal, it is possible to control the direction and the width of the beam emitted from the transmitter.
Meanwhile, a receiver 402 has a reversed configuration with respect to the transmitter 401. Signals received by a receiving antenna array composed of respective antenna elements 411-1-411-N are adjusted in both of amplitudes and phases or either ones of them in AWV control circuits 410-1-419-N, and they are combined. Then, received data decoded from the combined signal is externally output through a receiver circuit 409. As in the case of the transmitter 401, a process/arithmetic circuit 406 controls both of the amplitude and phase or either one of them for each of the AWV control circuits 410-1-419-N.
FIG. 5 is a conceptual diagram of a wireless communication system composed of two transmitting/receiving devices (400 and 500) each having the configuration shown in FIG. 4. The transmitting/receiving device 500 has K transmitting antennas and L receiving antennas.
Characteristics of a propagation path between two communication devices are expressed by a channel state information (CSI) matrix. It has been known that if this CSI matrix is determined, the optimal phase combination of the antenna array of the transmitting/receiving device can be obtained by using the SVD (Singular-Value Decomposition). However on the other hand, since the SVD is complex and requires a long processing time, it is very difficult to implement it for a transmission apparatus in which high-rate processing is required.
Accordingly, Document 4, for example, discloses a method for obtaining an optimal AWV with which the signal strength is maximized by adding a unitary matrix (e.g., Hadamard matrix) as phases of the antenna array and repeating the training of the antenna array of the transmitter and the training of the antenna array of the receiver. Although this method can reduce the processing time in comparison to the SVD, it still requires a certain time to obtain the optimal AWV combination since the method carries out the switching between the transmission and the reception repeatedly.
Meanwhile, Document 13 discloses a technique to optimize a transmitting/receiving beam direction by gradually increasing the beam resolution. However, this technique also requires measuring communication quality for a number of combinations of the transmitting/receiving beam directions by repeatedly carrying out the switching between the transmission and the reception, and thereby requiring a huge amount of time to obtain an optimal beam combination.
Furthermore, this document also brings up an idea called “quasi-omni (pseudo-nondirectional) pattern” as a beam having the lowest resolution. This quasi-omni pattern means a pattern having a substantially constant antenna gain over a very wide angle in the space around the transmitting/receiving device, though it is not a complete omni (nondirectional) pattern. Since it is often very difficult to obtain a complete omni pattern with millimeter-wave antenna arrays, this quasi-omni pattern is often used as a substitute in such cases.
When a link is to be established at the initial stage, it would be acceptable if the acquisition of an optimal AWV combination requires a long time. However, in a case where a link needs to be re-established when disconnection of the transmission occurs in the previously-established link, it is necessary to search for another optimal AWV combination in a short time. Furthermore, in the case of multipoint communication, a speedy search for an optimal AWV combination is also required because it requires the re-establishment of a plurality of links.